Signal receiver using data bit search in alternating time segments

ABSTRACT

A GNSS receiver and method using alternating “A” and “B” time segments for a reception time length of two or more data bits. The GNSS signal in an “A” time period comprising the “A” time segments is integrated for determining “A” magnitudes corresponding to code phase increments and the GNSS signal in a “B” time period comprising the “B” time segments is integrated for determining “B” magnitudes corresponding to code phase increments. A trial-and-error data bit search is performed for depolarizing data bit senses. The code phase increment showing the largest correlation level is used for acquisition of the GNSS signal and/or determination of the location where the GNSS is being received.

BACKGROUND OF THE INVENTION

1. Field of the Invention

The invention relates generally to a global positioning system (GPS)receiver and more particularly for a GPS receiver integrating a GPSsignal for an “A” time period having “A” time segments and integratingthe GPS signal for a “B” time period having “B” time segments, where the“A” time segments and the “B” time segments alternate, for signalacquisition at a low signal strength.

2. Description of the Background Art

The global positioning system (GPS) is a system using GPS satellites forbroadcasting GPS signals having information for determining location andtime. Each GPS satellite broadcasts a GPS signal having 20 milliseconds(ms) GPS data bits modulated by a 1 ms pseudorandom noise (PRN) codehaving 1023 bits or chips. The PRN code for each GPS satellite isdistinct, thereby enabling a GPS receiver to distinguish the GPS signalfrom one GPS satellite from the GPS signal from another GPS satellite.The 20 ms GPS data bits are organized into a frames of fifteen hundredbits. Each frame is subdivided into five subframes of three hundred bitseach.

Typically, when the GPS receiver is first turned on, it knows its ownapproximate location, an approximate clock time, and almanac orephemeris information for the locations-in-space of the GPS satellitesas a function of clock time. The GPS receiver processes the approximatetime, its approximate location, and the almanac or ephemeris informationto determine which of the GPS satellites should be in-view and generatesone or more GPS replica signals having carrier frequencies and apseudorandom noise (PRN) codes matching the estimated Dopplerfrequencies and the PRN codes of one or more of the in-view GPSsatellites. The GPS receiver correlates the carrier frequency, the PRNcode, and the PRN code phase of the incoming GPS signal to the replicasignals and then accumulates a correlation level. The process ofcorrelation and accumulation may need to be repeated many times until acorrelation level is found that exceeds a correlation threshold, therebyindicating GPS signal acquisition for the frequency, code, and codephase of the replica signals.

The incoming GPS signal has a low signal-to-noise ratio because of thespreading effect of the PRN code. The effect of the correlation andaccumulation process for despreading 1 ms or an epoch of the spread GPSsignal is to increase the signal-to-noise in order to be able torecognize the GPS data bits. This increase in signal-to-noise thatresults from the despreading is termed processing gain. Additionalprocessing gain can sometimes be achieved by correlating andaccumulating several epochs of the PRN code.

When signal acquisition is achieved the GPS receiver monitors the GPSdata bits until a hand over word (HOW) at the start of the subframe isrecognized. When the HOW is recognized, the GPS receiver reads time ofweek (TOW) in the GPS data bits in the HOW to learn a GPS-based clocktime. A current precise location-in-space of the GPS satellite is thencalculated from the GPS-based clock time and the ephemeris information.The code phase of the GPS replica signal is then used to calculate apseudorange between the location of the GPS receiver and thelocation-in-space of the GPS satellite. Typically, the ephemerisinformation is retained in memory in the GPS receiver from a previousoperational mode or is determined by reading additional GPS data bits.The geographical location fix is derived by linearizing the pseudorangeabout the range between the location-in-space of the GPS satellite andthe approximate location of the GPS receiver and then solving four ormore simultaneous equations having the locations-in-space and thelinearized pseudoranges for four or more GPS satellites.

The global positioning system is commonly used for determininggeographical location and/or time in commercial applications includingnavigation, timing, mapping, surveying, machine and agriculturalcontrol, vehicle tracking, and marking locations and time of events.Given such wide commercial application, it is clear that GPS receiversprovide a good value for many users. However, the global positioningsystem has been limited in several potential applications becauseexisting GPS receivers are unable to acquire a GPS signal unless the GPSsignal has a relatively clear line of sight to the GPS satellitesensuring strong GPS signals. Typically, this is not a problem where theGPS receiver is mounted on a platform such as a ship, airplane, farmtractor, or a vehicle traveling on an open highway. However, the signalstrength requirements of GPS receivers make it difficult to use GPSindoors or where the GPS signal may be weak due to the attenuation ofpassing through buildings or trees.

In order to increase the strength of the GPS signal within the GPSreceiver, workers in the art use techniques for increasing theprocessing gain above the standard processing gain that occurs bydespreading a single epoch of the 1 ms PRN code. For example, theadditional processing gain for integrating (correlating andaccumulating) ten coherent epochs is 10 log₁₀=10 decibels (dB) and theincreased processing gain for one-hundred coherent epochs is 10log₁₀100=20 decibels (dB). It would seem that one should increase thenumber of despread epochs indefinitely until enough processing gain isachieved for overcoming the GPS signal attenuation caused by buildingsand trees. Unfortunately, every 20 ms the C/A PRN code may be invertedwith a new GPS data bit, thereby nullifying the processing gain forintegration times beyond 20 ms. Accordingly, there continues to be aneed for improvements in GPS receivers and methods for acquisition ofweak GPS signals.

SUMMARY OF THE INVENTION

In order to more easily follow the summaries of first and secondembodiments, the reader may first want to refer to FIGS. 12A and 12B.

A first embodiment is a method for receiving an incoming signal havingdata bits spread by a spreading code, comprising: integrating theincoming signal at code phase increments of the spreading code with twoor more data bit search patterns in an “A” integration time periodcomprising at least two non-contiguous “A” time segments for determiningmagnitudes corresponding to the code phase increments, respectively, foreach of the search patterns; integrating the incoming signal at the codephase increments with the search patterns for a “B” integration timeperiod comprising “B” time segments alternating with the “A” timesegments for determining magnitudes corresponding to the code phaseincrements, respectively, for each of the search patterns; and using alargest of the magnitudes for determining a particular one of the codephase increments for receiving the incoming signal.

A second embodiment is a receiver for acquiring an incoming signalhaving data bits spread by a spreading code, comprising: a correlationmachine to integrate the incoming signal at code phase increments of thespreading code with two or more data bit search patterns in an “A”integration time period comprising at least two non-contiguous “A” timesegments for determining magnitudes corresponding to the code phaseincrements, respectively, for each of the search patterns; thecorrelation machine further to integrate the incoming signal at the codephase increments with the search patterns for a “B” integration timeperiod comprising “B” time segments alternating with the “A” timesegments for determining magnitudes corresponding to the code phaseincrements, respectively, for each of the search patterns; and anavigation signal processor configured to use a largest of themagnitudes for determining a particular one of the code phase incrementsfor receiving the incoming signal.

Systems that use the first and second embodiments are illustrated inFIGS. 10, 11A-B, 12A-B, 13, 14 and 15 and described in detail in theaccompanying detailed textual descriptions. Further descriptive materialis illustrated in FIGS. 1A-B, 2, 3, 4, 5, 6A-F, 7A-I, 8 and 9 anddescribed in detail in the accompanying detailed descriptions.

Briefly, this descriptive material shows a way to acquire a signalhaving data bits spread by a spreading code even when the spreadspectrum signal is weak by integrating the signal in separateinterleaved time periods termed “A” and “B” time periods where the “A”time period includes “A” time segments and the “B” time period includes“B” time segments that alternate with the “A” time segments. Knownpolarities of expected data bits having a known expected reception timeperiod are used to invert or not invert (depolarize) the signalseparately for each time segment. By constructing the time segments tobe one-half a data bit period, at least one of the “A” and “B” timesegments avoids transitions in the GPS data bits, thereby enabling acontinuous integration without the nullifying effect of inversions ofthe data bits. An embodiment is described in terms of a spread spectrumglobal positioning system (GPS) signal. In order to more easily followthe summaries of embodiments, the reader may first want to refer toFIGS. 6A-E.

This material further shows the polarities of certain 20 millisecond(ms) GPS data bits are known beforehand and their expected receptiontimes are known to within ±10 ms. “A” time segments of 10 ms and “B”time segments of 10 ms alternate. The GPS signal in each of the 10 ms“A” time segments is depolarized according to the known polarities ofthe expected GPS data bits. Likewise, the GPS signal each of the 10 ms“B” time segments is depolarized according to the known polarities ofthe same GPS data bits. After accounting for polarities of the expecteddata bits, the GPS signal during an “A” time period including all the 10ms “A” time segments is coherently integrated (accumulated) forproviding an “A” time period magnitude for each potential code phase ofa pseudorandom (PRN) spreading code of the GPS signal. Likewise, the GPSsignal during a “B” time period including all the 10 ms “B” timesegments is coherently integrated for providing a “B” time periodmagnitude for each potential GPS code phase. The strongest of the timeperiod magnitudes is detected and then compared to a correlationthreshold. When the threshold is exceeded, the GPS code phase thatyielded the strongest time period magnitude is used for GPS signalacquisition.

This material further shows “A” time period may also include “A”augmentation time segments corresponding to certain “B” time segmentswhen consecutive data bits have no change in polarity. Likewise, the “B”time period may also include certain “B” augmentation time segmentscorresponding to certain “A” time segments when consecutive data bitshave no change in polarity.

In a time domain version of this material, the GPS signal is integratedby accumulating correlation levels for the “A” time period and the “B”time period at each potential code phase for providing the “A” timeperiod magnitudes and the “B” time period magnitudes, respectively. In afrequency domain version of this material, the “A” time periodmagnitudes and the “B” time period magnitudes are integrated for eachpotential code phase using fast Fourier transform (convolution)techniques.

This material further shows each of the 10 ms “A” and “B” time segmentsincludes 10 epochs for the 1 ms GPS spreading code. An advantage of thisis that a large signal processing gain is achieved due to the length oftime of the “A” time period (or the “B” time period), thereby improvingthe probability of rapid acquisition of a weak signal.

This material further shows a number N of types of alternating timesegments, where the “A” and “B” alternating time segments represents anembodiment for N=2. For example for N=4 (FIG. 6F), “A”, “B”, “C”, and“D” time segments, each time segment having one-half a data bit timeperiod, alternate for an expected reception time period for a sequenceof known data bits. Increasing the number N of types of time segmentsallows a tradeoff that increases the tolerance for time error at theexpense of reducing the processing gain.

This material further shows a method for receiving an incoming signalhaving data bits spread by a spreading code, comprising: integrating twoof more data bit each results for the incoming signal in an “A” timeperiod comprising “A” time segments for determining “A” magnitude setscorresponding to the search results, respectively, the “A” magnitudesets comprising “A” magnitudes corresponding to code phase incrementsfor the incoming signal; integrating the data bit search results in a“B” time period comprising “B” time segments, the “B” time segmentsalternating with the “A” time segments, for determining “B” magnitudesets corresponding to the search results, respectively, the “B”magnitude sets comprising “B” magnitudes corresponding to code phaseincrements for the incoming signal; and using a strongest of the “A” and“B” magnitudes for determining a particular one of the code phaseincrements for receiving the incoming signal.

This material further shows a receiver for acquiring an incoming signalhaving data bits spread by a spreading code, comprising: a correlationmachine to integrate two of more data bit search results for theincoming signal in an “A” time period comprising “A” time segments todetermine “A” magnitude sets corresponding to the search results,respectively, the “A” magnitude sets comprising “A” magnitudescorresponding to code phase increments for the incoming signal; thecorrelation machine to integrate the data bit search results in a “B”time period comprising “B” time segments, the “B” time segmentsalternating with the “A” time segments, to determine “B” magnitude setscorresponding to the search results, respectively, the “B” magnitudesets comprising “B” magnitudes corresponding to code phase incrementsfor the incoming signal; and a navigation signal processor configured touse a strongest of the “A” and “B” magnitudes for determining aparticular one of the code phase increments for receiving the incomingsignal.

These and other embodiments of the present invention will no doubtbecome obvious to those of ordinary skill in the art after having readthe following detailed description of the embodiments which areillustrated in the various figures.

IN THE DRAWINGS

FIGS. 1A and 1B are flow charts of alternative methods for acquiringspread spectrum signals using “A” and “B” time periods;

FIG. 2 is a block diagram of a GPS receiver using “A” and “B” timeperiods;

FIG. 3 is a block diagram of a time domain correlation machine for a GPSreceiver;

FIG. 4 is a block diagram;

FIG. 5 is a block diagram of an AB processor for the correlationmachines of FIGS. 3 and 4;

FIGS. 6A and 6B are time charts showing “A” and “B” time segments for acontinuous expected reception time period and a discontinuous expectedreception time period, respectively;

FIG. 6C is a time chart showing “A” and “B” augmentation time segments;

FIGS. 6D and 6E are time charts for time errors of 5 and 10milliseconds, respectively, showing “A” and “B” time period magnitudes;

FIG. 6F is a time chart showing “A”, “B”, “C”, and “D” time segments;

FIGS. 7A-I are flow charts showing embodiments, respectively, havingfirst through ninth arrangements, respectively, for depolarizing,separating, and integrating an incoming signal;

FIG. 8 is a block diagram of an N processor for N types of time periods;

FIG. 9 is a block diagram of a GPS receiver using N types of timeperiods;

FIG. 10 is a block diagram of a GPS receiver using a data bit searchgenerator with “A” and “B” time periods;

FIGS. 11A and 11B are flow charts of a method for acquiring spreadspectrum signals using a data bit search with “A” and “B” time periods;

FIGS. 12A and 12B are time charts showing “A” and “B” time segments andshowing accumulated “A” and “B” magnitudes for a correct data bit searchpattern for a correct code phase increment for two and three data bittime lengths, respectively;

FIGS. 13 and 14 are block diagram of a time and frequency domaincorrelation machines with a data bit search generator; and

FIG. 15 is a flow chart of a data bit search method.

DETAILED DESCRIPTION

The FIGS. 1A and 1B are flow charts illustrating variations for fastacquisition of weak global positioning system (GPS) signals. At thestart, GPS-based time is known beforehand to within one-half the timeperiods of the GPS data bits; the polarities of a sequence of GPS databits are known beforehand (stored sequence 122, FIG. 2); and ephemerisinformation is known or obtained. A method of the present invention isused for acquiring the GPS signal for a first GPS satellite. Then, themethod of the present invention or conventional methods may be used foracquiring the GPS signal from other GPS satellites for determining alocation fix.

Referring to FIG. 1A, in a step 52 the incoming GPS signal is received.In a step 54 the GPS signal is downconverted. In a step 56 the start ofan expected reception time period for an incoming sequence of certainexpected GPS data bits is detected. The expected reception time periodmay be wider than the time period for the incoming sequence in order toaccommodate the time delays for differing distances to GPS satellites.In a step 58, the downconverted GPS signal is sampled and arepresentation of the GPS signal is stored during the expected receptiontime period. In a step 62 the expected reception time period associatedwith the stored signal representation is organized into alternating “A”time segments and “B” time segments as shown in FIGS. 6A-E and describedin the accompanying detailed descriptions below.

The stored signal representation is depolarized in a step 64 for thepolarities of the stored sequence (122, FIG. 2). In a step 66 anappropriate Doppler frequency is chosen and the depolarized signalrepresentation is integrated (coherently accumulated) for each potentialGPS code phase for an “A” time period including all the “A” timesegments. Similarly, in a step 68 the depolarized signal representationis integrated (coherently accumulated) using the Doppler frequency foreach potential GPS code phase for the “B” time period including all the“B” time segments. The signal integrations for the “A” time periodprovide “A” time period magnitudes corresponding to the potential codephases, respectively, and the signal integrations for the “B” timeperiod provide “B” time period magnitudes corresponding to the samepotential code phases, respectively. Then, in a step 70 the code phaseincrement that results in the strongest of the “A” time periodmagnitudes, the “B” time period magnitudes, or combination of the “A”:and “B” time period magnitudes at the same code phase increment istested against a correlation threshold and used for signal acquisitionwhen the correlation threshold is exceeded. In order to find a timeperiod magnitude that exceeds the correlation threshold the steps 66,68, and 70 may be iterated using different assumptions for carrierfrequency.

Referring to FIG. 1B, the steps 52-62 are described above in thedetailed description of FIG. 1A. Then, in a step 72 the stored signalrepresentation is integrated for the “A” time segments for providing “A”time segment integrations. Similarly, in a step 74 the stored signalrepresentation is integrated for the “B” time segments for providing “B”time segment integrations.

The “A” time segment integrations are depolarized in a step 76 for thepolarities of a stored sequence (122 in FIG. 2) representing theincoming sequence of certain expected GPS data bits for the “A” timesegments of the expected reception time period. Similarly, in a step 78the “B” time segment integrations are depolarized for the polarities ofthe stored sequence (122 in FIG. 2) for the “B” time segments of theexpected reception time period.

The “A” depolarized integrations are integrated (coherently accumulated)in a step 82 for the “A” time period including all the “A” time segmentsfor providing the “A” time period magnitudes. Similarly, in a step 84the “B” depolarized integrations are integrated (coherently accumulated)for the “B” time period including all the “A” time segments forproviding the “B” time period magnitudes. The signal integrations forthe “A” time period provide “A” time period magnitudes corresponding tothe potential code phases, respectively, and the signal integrations forthe “B” time period provide “B” time period magnitudes corresponding tothe same potential code phases, respectively. Then, in the step 70described above, the code phase increment that results in the strongestof the “A” time period magnitudes, the “B” time period magnitudes, or acombination of the “A” and “B” time period magnitudes is tested againsta correlation threshold and used for signal acquisition and/or trackingwhen a correlation threshold is exceeded. In order to find a time periodmagnitude that exceeds the correlation threshold the steps 72-84 and 70may be iterated using different assumptions for carrier frequency.

The FIGS. 1A and 1B and the accompanying descriptions illustrate anddescribe variations in the order of the steps for depolarizing andintegrating the representations of the GPS signal. The GPS signal isintegrated in one or more steps for the 10 ms “A” and the 10 ms “B” timesegments and then the “A” and “B” time segment integrations areintegrated (coherently accumulated) for the “A” and “B” time periods.The step of depolarizing may be applied to a representation of the GPSsignal before the 10 ms integrations, within stages of the 10 msintegrations of the individual “A” and “B” time segments, to a GPSreplica signal used in the 10 ms integrations, or to the “A” and “B”time segments before the 10 ms integrations are integrated (coherentlyaccumulated) for the “A” and “B” time periods. Also, the separation ofthe processing for the “A” and “B” time segments may be made at anypoint before or after the depolarization up to the integration of the“A” time segments into the “A” time period and the integration of the“B” time segments into the “B” time period.

FIG. 2 is a block diagram of a signal receiver referred to by a generalreference number 100. The receiver 100 is described in terms of a globalpositioning system (GPS) receiver for receiving incomingcoarse/acquisition (C/A) GPS signals. However, those of ordinary skillthe art of spread spectrum radio receivers will see that the blockdiagram illustrated in FIG. 2 and described below can be applied forreceiving other types of signals, especially for receiving other typesof direct fast sequence spread spectrum signals including precision P orP(Y) code GPS signals.

The GPS receiver 100 includes a GPS antenna 102, a frequencydownconverter 104, a sampler 106, a stored signal memory 108, acorrelation machine 112, a microprocessor 114, and a program memory 116.The GPS antenna 102 converts incoming airwave GPS signals from GPSsatellites or pseudolites to conducted GPS signals and passes theconducted GPS signals to the frequency downconverter 104. The frequencydownconverter 104 includes a reference generator 105, local oscillatorsderiving their frequencies from the reference generator 105, and mixersfor downconverting the GPS signals to a lower frequency and passing thelower frequency GPS signals to the sampler 106. The sampler 106 samplesthe downconverted GPS signals for providing in-phase (I) and quadraturephase (Q) samples. The signal memory 108 stores the I and Q samples fora selected time period as a stored signal representation for later useby the correlation machine 112. In an alternative embodiment thecorrelation machine 112 processes a representation of the incomingsignal in real time as it arrives.

The correlation machine 112 integrates the signal representation forproviding correlation magnitudes corresponding to code phases,respectively, of the pseudorandom noise (PRN) spreading code in the GPSsignal. Typically, the correlation machine 112 has several independentchannels where each channel processes the stored signal representationthe GPS signal for one GPS satellite at a time. For example, a 12channel correlation machine 112 could be aligned for acquiring and/ortracking GPS signals from 12 GPS satellites, respectively, or eachchannel of the correlation machine 112 could be aligned for sharing thetask of acquiring and/or tracking the GPS signal from one GPS satelliteor several channels could be aligned for acquiring the GPS signal fromseveral GPS satellites while several channels are aligned for acquiringand/or tracking the GPS signal from one GPS satellite. The expectedreception time period for the stored signal representation for each GPSsatellite is offset in time several milliseconds depending upon thelocation of the GPS satellite transmitting the signal.

The microprocessor 114 reads programmed instructions in the programmemory 116 for controlling the elements of the GPS receiver 100. Theprogram memory 116 includes a stored sequence 122, an acquisitiondetector 124, and a signal and navigation program 126. The storedsequence 122 includes data for known polarities of certain expected GPSdata bits having respective expected reception times. In an embodiment,the stored sequence 122 includes expected GPS data bits forming acontinuous sequence for a continuous expected reception time period.However, the stored sequence 122 can be any two GPS data bits, eithercontiguous or separated, having known polarities and expected receptiontimes. The acquisition detector 124 includes instructions for using thecorrelation magnitudes from the correlation machine 112 for determiningthe code phase of the PRN spreading code to use for GPS signalacquisition. The signal and navigation program 126 includes instructionsfor directing the microprocessor 114 for signal acquisition, signaltracking, location and time fixes, and control for the functions of theGPS receiver 100.

The GPS receiver 100 also includes an expected sequence timer 132, andan AB timer 134. The sequence timer 132 uses the reference generator 105for maintaining GPS time to within ±10 ms for gating the sampler 106and/or the signal memory 108 for storing the sampled downconverted GPSsignal during the expected reception time period. There are several waysin which GPS time can be maintained to within ±10 ms. For relativelyshort periods of time, for example a few hours to a few days, anaccurate GPS-based time can be maintained with an accurate temperaturecontrolled clock. For longer periods of time, an accurate GPS-time canbe received in a time signal transmitted from a time source. Thesequence timer 132 also triggers the AB timer 134 for generating AB timesegment signals for organizing the expected reception time period intoalternating 10 ms “A” time segments and 10 ms “B” time segments asillustrated in FIGS. 6A-E and described in the accompanying detaileddescription below. Typically, the “A” and “B” time segments for one GPSsatellite are offset in time from the “A” and “B” time segments foranother GPS satellite in order to accommodate different distances to thesatellites.

The correlation machine 112 includes an AB processor 136. The ABprocessor 136 provides “A” time period magnitudes for an “A” time periodhaving accumulated integrations for the “A” time segments (and “A”augmentation time segments described below) and provides “B” time periodmagnitudes for a “B” time period having accumulated integrations for the“B” time segments (and “B” augmentation time segments described below).The correlation machine 112 including the AB processor 136 integratesthe stored signal representation of the GPS signal during the “A” timeperiod for providing an “A” time period magnitude for each potential GPScode phase and during the “B” time period for providing a “B” timeperiod magnitude for each potential GPS code phase.

A replica generator 138 preferably included in the correlation machine112 generates GPS replica signals for the carrier frequency and thespreading code of the GPS signals as represented by the stored signalrepresentation of the GPS signals. The C/A code of the GPS signal has a1023 bit or chip spreading code. The replica generator 138 issues areplica signal for all 1023 chips either in the time or frequency domainfor both I and Q in one-half chip or smaller increments of code phase.The correlation machine 112 in an embodiment provides I and Qcorrelation levels in increments that are slightly less than one-halfchip. Preferably about 2048 code phase increments are provided for I andabout 2048 code phase increments are provided for Q to the AB processor136. During the “A” time period, the AB processor 136 accumulates andcombines the I and Q levels for providing the “A” time periodmagnitudes; and during the “B” time period, the AB processor 136accumulates and combines the I and Q levels for providing the “B” timeperiod magnitudes. The correlation machine 112 and integration processare illustrated and described in greater detail in FIGS. 3-4 and theaccompanying descriptions below.

The “A” time period magnitudes and the “B” time period magnitudes fromthe correlation machine 112 are processed by the microprocessor 114using instructions in the acquisition detector 124 for determining theincrement of code phase for the replica code that appears to match thecode phase of the incoming GPS signal. This replica code phase is thenprocessed for GPS signal acquisition by the microprocessor 114 accordingto instructions in the signal and navigation program 126.

There are several embodiments for the acquisition detector 124 that canprocess the “A” and “B” time period magnitudes in order to find the codephase for acquisition. In an embodiment, the strongest of the timeperiod magnitudes, whether one of the “A” time period magnitudes or the“B” time period magnitudes, is tested against a correlation threshold.In another embodiment, the “A” time period magnitude and the “B” timeperiod magnitude for a code phase are combined (incoherently) or theirsquares are added to form an AB time period magnitude for that codephase. When the strongest combined time period magnitude exceeds thecorrelation threshold, the code phase that resulted in that combinedtime period magnitude is used for signal acquisition and tracking. Oncethe code phase for acquisition of the GPS signal from a single GPSsatellite is determined, conventional methods can be used for acquiringthe GPS signal from other GPS satellites. When the strongest magnitudedoes not exceed the required correlation threshold, then a differentDoppler frequency, a different PRN code, a different GPS time, or thelike is tried and the process is repeated; or the GPS receiver 100 maygo into a standby mode to conserve power.

The stored sequence 122 may be a fixed sequence stored in the programmemory 116. However, typically the stored sequence 122 is calculateddepending upon recent information and then stored in the program memory116 after calculation. One sequence of GPS data bits that can be usedfor the stored sequence 122 is the time-of-week (TOW) in thehand-over-word (HOW) that is broadcast in each subframe of the GPSsignal. The TOW is the 17 most significant bits (MSB)s corresponding toa TOW-count at the epoch which occurs at the leading edge of the nextfollowing subframe. The polarities of this sequence of 17 bits are knownby knowing the GPS-based time which is maintained by the GPS receiver100 to within ±10 ms for use by the expected sequence timer 132. The 17bits of the TOW result in a total expected reception time period ofabout 340 ms. Extra time may be required time error or to account forearly and late edges times. Those skilled in the art will be able todetermine other sequences within the GPS subframes that can be used. Amore complete understanding of the HOW and TOW and other sequenceswithin the GPS signal is available in published form from NavtechSeminars & Navtech Bookstore and Software Store, of Arlington, Va.,under title of GPS Interface Control Document ICD-GPS-200 which isincorporated herein by reference.

The expected GPS data bits do not need to be continuous or consecutive.For non-consecutive data bits, the sequence timer 132 provides triggersto the AB timer 134 for the expected reception time period for eachsection of the stored sequence 122. However, it should be understoodthat a wider separation between the beginning of the first time segmentand the end of the last time segment results in a narrower carrierfrequency range of the GPS signal for the correlation process. When thecarrier frequency range is narrow, the integration process may need tobe iterated for many carrier center frequencies before a satisfactorycode phase is found for signal acquisition. Preferably, the referencegenerator 105 generates a reference signal having a stable frequency anda capability of being updated or stabilized further with a frequencystandard signal transmitted from a frequency standard source.

Using more than two types for sequential 10 ms time segments reduces therequirement for accurate time but also reduces the signal processinggain. Doubling the number of types of time segments halves the timeaccuracy requirement. For example, four types of alternating 10 ms timesegments (“A” then “B” then “C” then “D” then “A” again and so on) couldbe used for reducing the time accuracy requirement to ±20 ms but alsoreduces the total time for any one of the “A”, “B”, “C”, and “D” timeperiods by a factor of two, thereby reducing the signal processing gain.A time chart illustrating “A”, “B”, “C”, and “D” alternating timesegments is shown in FIG. 6F and described in the accompanying detaileddescription.

FIG. 3 is a block diagram showing the signal memory 108 and a timedomain correlation machine 112A that is an embodiment of the correlationmachine 112 described above. The block diagram of the correlationmachine 112A shows depolarizers 143-149 in optional placements withinthe correlation machine 112A. One of the depolarizers 143-149depolarizes a representation of the GPS signal by inverting the GPSsignal representation for one polarity for a GPS data bit in the storedsequence 122 and not inverting the GPS signal representation for theother polarity. Only one of the depolarizers 143-149 at only one of theplacements is used in any one embodiment.

The block diagram of the correlation machine 112A illustrates a singlechannel for determining “A” and “B” time period magnitudes forrespective code phases. However, an embodiment of the GPS receiver 100will include several such channels in the correlation machine 112A whereeach channel operates effectively in parallel.

The correlation machine 112A includes a replica generator 138A that is aversion of the replica generator 138 described above. The replicagenerator 138A includes a carrier numerically controlled oscillator(NCO) 152, a code numerically controlled oscillator (NCO) 154, apseudorandom (PRN) coder 156, and a code phase shifter 158. The carrierNCO 152, code NCO 154, PRN coder 156, and code phase shifter 158 arecontrolled by the microprocessor 114. The carrier NCO 154 issues I and Qreplica carrier signals for the expected carrier frequency of thedownconverted GPS signal. The code NCO 154 issues a code rate signal forthe expected code rate of the GPS signal to the PRN coder 156. The PRNcoder 156 generates a selected PRN code at the repetition rate of thecode rate signal. The selected PRN code may be selected to be differentor the same for each of the channels of the correlation machine 112Adepending upon the status of the GPS receiver 100. The code phaseshifter 158 issues I and Q replica versions of the PRN code in sampleincrements of code phase of preferably one-half chip at a controlledcode phase. In order to determine the correct code phase for signalacquisition, the code phase shifter 158 shifts the PRN code in shiftincrements of code phase that are one-half chip or smaller, preferablyone-half chip. Alternatively, the code phase shifter 158 is not requiredif the PRN coder 156 issues I and Q replica code signals for incrementalcode phase shifts in parallel.

The correlator machine 112A also includes I and Q carrier multipliers162, and I and Q code multipliers 164. Although only one set of carriermultipliers 162 and one set of code multipliers 164 are shown, severalsets may be included for faster parallel processing. Multiple codemultipliers 164 are required where the replica code signal includes thecode phase shifts in parallel. The correlation machine 112A alsoincludes I and Q accumulators 166 and the AB processor 136 (or an Nprocessor 800, FIGS. 8-9). The processing of the I samples from thesignal memory 108 will now be described with the understanding that theQ samples are processed in the same way. The I carrier multiplier 162correlates I samples from the signal memory 108 with the I samples ofthe replica carrier signal from the carrier NCO 152 for providing an Ibaseband (or pseudo-band) representation of the GPS signal. The I codemultiplier 164 correlates the I baseband signal with the I replica PRNcode from the code phase shifter 158 for providing I code correlationsto the I accumulator 166 for each increment of code phase at the currentincrement of replica code phase shift.

The I accumulator 166 despreads the GPS signal by accumulating(integrating) the I code correlations for the current increment ofreplica code phase shift for a selected time period, preferably eitherone complete code time period at 1 ms for ten times or ten complete codeperiods at 10 ms for one time. For the example of slightly less thanone-half chip incremental code phase shift for the 1023 chip C/A PRNcode, for either 1 ms or 10 ms correlation time periods, the Iaccumulator provides preferably about 2048 I integrations. Theaccumulated I integrations are then passed to the AB processor 136 (oran N processor 800, FIGS. 8-9). The same process is followed for the QGPS signal representations for providing accumulated Q integrations tothe AB processor 136 (or N processor 800, FIGS. 8-9).

The depolarized replica signal is used within the correlation machine112A for depolarizing a representation of the incoming signal. Theplacement of the depolarizer 143 shows an optional placement fordepolarizing the I and Q replica carrier signal from the carrier NCO 152for providing a depolarized replica signal. The placement of thedepolarizer 144 shows an optional placement for depolarizing the PRNcode from the PRN coder 156 for providing a depolarized replica signal.The placement of the depolarizer 145 shows an optional placement fordepolarizing the I and Q PRN codes from the code phase shifter 158 forproviding a depolarized replica signal. The placement of the depolarizer146 shows an optional placement for depolarizing the I and Q storedsignal representation from the signal memory 108. The placement of thedepolarizer 147 shows an optional placement for depolarizing the I and Qbaseband GPS signal from the carrier multipliers 162. The placement ofthe depolarizer 148 shows an optional placement for depolarizing the Iand Q correlations from the code multipliers 164. The placement of thedepolarizer 149 shows an optional placement for depolarizing the I and Qintegrations from the accumulator 168.

FIG. 4 is a block diagram showing the signal memory 108 and a frequencydomain correlation machine 112B that is an embodiment of the correlationmachine 112 described above. The block diagram of the correlationmachine 112B shows the depolarizers 143-144 and 146-147 described above,and a depolarizer 172 in optional placements within the correlationmachine 112B. One of the depolarizers 143-144, 146-147, or 172depolarizes a representation of the GPS signal by inverting the GPSsignal representation for one polarity for a GPS data bit in the storedsequence 122 and not inverting the GPS signal representation for theother polarity. Only one of the depolarizers 143-144, 146-147, and 172at only one of the placements is used in any one embodiment.

The block diagram of the correlation machine 112B illustrates a singlechannel for determining “A” and “B” time period magnitudes forrespective code phases. However, an embodiment of the GPS receiver 100will include several such channels in the correlation machine 112B whereeach channel operates effectively in parallel.

The correlation machine 112B includes a replica generator 138B that is aversion of the replica generator 138 described above. The replicagenerator 138B includes the carrier NCO 152, the code NCO 154, and thePRN coder 156 as described above, and a replica fast Fourier transformer174. The replica fast Fourier transformer (FFT) 178 performs a complexfast frequency transform on the PRN code from the PRN coder 156 forproviding frequency domain I and Q replica code signals corresponding toincrements of code phase of preferably one-half chip.

The correlator machine 112B also includes the I and Q carriermultipliers 162 described above, a signal fast Fourier transformer (FFT)176, I and Q code multipliers 178, an inverse fast Fourier transformer(IFFT) 180, and the AB processor 136 (or N processor 800, FIGS. 8-9).The signal FFT 176 performs a complex fast Fourier transform on the Iand Q baseband signals from the I and Q carrier multipliers 162 forproviding frequency domain I and Q baseband signals. The I codemultiplier 178 multiplies the frequency domain I baseband signal by thefrequency domain I replica code signal for providing a frequency domainI despread GPS signal to the IFFT 180. Similarly, the Q code multiplier178 multiplies the frequency domain Q baseband signal by the frequencydomain Q replica code signal for providing a frequency domain Q despreadGPS signal to the IFFT 180. The IFFT 180 performs a complex inverse fastFourier transform on the frequency domain I and Q despread GPS signalsfor providing I and Q correlation levels for all potential code phaseincrements in parallel for a selected time period of preferably 10 ms.For an example of slightly less than one-half chip increments of replicacode, for the 1023 chip C/A PRN code, the IFFT 180 preferably providesabout 2048 I integrations and about 2048 Q integrations. The I and Qcorrelation levels are then passed to the AB processor 136 (or Nprocessor 800, FIGS. 8-9).

The placement of the depolarizer 172 shows an optional placement fordepolarizing the I and Q integrations after the IFFT 180. Thedepolarizers 143-144 and 146-147 are optionally placed within thecorrelation machine 112B as described above in the detailed descriptionaccompanying FIG. 3 for the correlation machine 112A.

FIG. 5 is a block diagram of the AB processor 136 for receiving I and Qintegrations from the I and Q accumulators 166 of the time domaincorrelation machine 112A or the IFFT 180 of the frequency domaincorrelation machine 112B. One of the depolarizers 143-149 or 172 is inplace so that the I and Q integrations are depolarized according to thepolarities in the stored sequence 122.

The AB processor 136 includes an AB switch 202, accumulators 206-209, an“A” combiner 212, and a “B” combiner 214. The AB switch 202 separates arepresentation of the incoming signal into the “A” time period and the“B” time period. The “A” time period includes “A” time segments and “A”augmentation time segments (FIG. 6A-E). The “B” time period includes “B”time segments and “B” augmentation time segments (FIG. 6A-E). There areseveral variations (FIGS. 7A-I) regarding the placement of the AB switch202 within the GPS receiver 100 for separating the incoming signalaccording to the “A” and “B” time periods. In one variation the ABswitch 202 separates the stored signal representation where the storedsignal representation enters the correlation machine 112 from the signalmemory 108. In this variation the correlation machine 112 integrates thestored signal representation for the “A” time period separately from thestored signal representation for the “B” time period. In anothervariation the AB switch 202 is disposed within the correlation machine112 so that signal can be processed in time periods of up to the 10 mstime periods of the “A” and “B” time segments before the AB switch 202.

In an embodiment, the AB switch 202 includes an I AB switch 202I and a QAB switch 202Q. The I AB switch 202I receives the depolarized Iintegrations from the I accumulator 166 for the time domain correlationmachine 112A or the IFFT 180 for the frequency domain correlationmachine 112B. Similarly, the Q AB switch 202Q receives the depolarized Qintegrations from the Q accumulator 166 for the time domain correlationmachine 112A or the IFFT 180 for the frequency domain correlationmachine 112B. In FIG. 5, the I AB switch 202I and the Q AB switch 202Qare shown for the “A” time period.

For the “A” time period, the I AB switch 202I passes the depolarized Iintegrations to the accumulator 206. For the “B” time period, the I ABswitch 202I passes the depolarized I integrations to the accumulator207. The accumulator 206 accumulates the depolarized I integrations forthe “A” time period for providing an “A” I magnitude accumulation foreach code phase. The accumulator 207 accumulates the depolarized Iintegrations for the “B” time period for providing a “B” I magnitudeaccumulation for each code phase.

The Q AB switch 202Q for the “A” time period passes the depolarized Qintegrations to the accumulator 208. For the “B” time period the Q ABswitch 202Q passes the depolarized Q integrations to the accumulator209. The accumulator 208 accumulates the depolarized Q integrations forthe “A” time period for providing an “A” Q magnitude accumulation foreach code phase. The accumulator 209 accumulates the depolarized Qintegrations for the “B” time period for providing a “B” Q magnitudeaccumulation for each code phase.

The A combiner 212 adds the square of the “A” I magnitude accumulationfrom the accumulator 206 to the square of the “A” Q magnitudeaccumulation from the accumulator 208 for each code phase increment forproviding the “A” time period magnitudes. The B combiner 214 adds thesquare of the “B” I magnitude accumulation from the accumulator 207 tothe square of the “B” Q magnitude accumulation from the accumulator 209for each code phase increment for providing the “B” time periodmagnitudes. At this point for code phase increments of slightly lessthan one-half chip there will be preferably about 2048 “A” time periodmagnitudes and about 2048 “B” time period magnitudes. As describedabove, the code phase increment that results in the strongest of the “A”time period magnitudes, the “B” time period magnitudes, or combinationof the “A” and “B” time period magnitudes is tested against acorrelation threshold and used for signal acquisition when thecorrelation threshold is exceeded.

As described above the “A” I and Q magnitude accumulations and “B” I andQ magnitude accumulations are preferably linear functions proportionalto the levels for the I and Q integrations of the GPS signal, whereasthe “A” and “B” time period magnitudes are preferably proportional tothe sum of the squares the I and Q magnitude accumulations (I²+Q²).However, it should be noted that the “A” and “B” time period magnitudescan be some other convenient non-decreasing function of the “A” and “B”I and Q magnitude accumulations, such as the sum (I+Q) or square root ofthe sum of the squares (√(I²+Q²)).

The “A” and “B” time segments can be processed separately through thecorrelation machines 112A and 112B. In a variation of the correlationmachine 112A, the I carrier multiplier 162, the I code multiplier 164,and the I accumulator 166 integrate representations of the GPS signalfor “A” time segments and separately for “B” time segments for providingdepolarized I integrations for “A” time segments and depolarized Iintegrations for “B” time segments, respectively, and likewise for Q.The AB processor 136 then uses the accumulators 206-209 and thecombiners 212 and 214 as described above. Similarly, for the correlationmachine 112B, the carrier multipliers 162, the signal FFT 176, the codemultipliers 178, and the IFFT 180 integrate representations of the GPSsignal for “A” time segments and separately for “B” time segments forproviding depolarized I integrations for “A” time segments anddepolarized I integrations for “B” time segments, respectively, andlikewise for Q.

FIGS. 6A-B illustrate the “A” and “B” time segments with respect toexemplary sequences of GPS data bits. Each of the “A” time segments andeach of the “B” time segments is preferably about one-half the timelength of a data bit. For the exemplary case of C/A GPS, the data bitshave a time period of 20 ms, thereby the “A” and “B” time segmentspreferably have periods of about 10 ms. The “A” time segments start atthe start of expected reception time period of the first bit of a knownsequence of expected GPS data bits (t=0 ms) for a case where GPS-basedtime is accurate to within a small portion of a millisecond within theGPS receiver 100. The “A” time segments and then repeat each 20 msthereafter for the expected reception time period. The “B” time segmentsstart 10 ms later (t=10 ms) and repeat 20 ms thereafter for the expectedreception time period in order to alternate with the “A” time segments.FIG. 6A shows the expected reception time period for a continuoussequence of expected GPS data bits. FIG. 6B shows the expected receptiontime period for a discontinuous sequence of expected GPS data bits. The“A” time period includes only those “A” time segments for the expectedGPS data bits while leaving a gap for the time period between theexpected GPS data bits and likewise the “B” time period includes onlythose “B” time segments for the expected GPS data bits while leaving agap for the time period between the expected GPS data bits.

FIG. 6C shows the “A” and “B” time segments numbered A1-11 and B1-11,respectively, in positions with respect to the sequence of expected databits for a locally maintained GPS-based time that is exactly accurate.For an error in the GPS-based time of up to plus or minus 10 ms, it canbe seen that the B4, the B8 , and the B9 time segments denoted withasterisks (*), termed herein “A” augmentation time segments, correspondto the same polarity in the sequence as they have for the exactlycorrect time. Accordingly, in order to increase processing gain of the“A” time period magnitudes, the “A” time period can be constructed toinclude not only the A1-A11 time segments but also the “A” augmentationtime segments B4*, B8*, and B9*. Similarly, for an error in theGPS-based time of up to plus or minus 10 ms, it can be seen that the A5,the A9, and the A10 time segments denoted with asterisks (*), termedherein “B” augmentation time segments, correspond to the same polarityin the sequence as they have for the exactly correct time. Accordingly,in order to increase processing gain of the “B” time period magnitudes,the “B” time period can be constructed to include not only the B1-B11time segments but also the “B” augmentation time segments A5*, A9*, andA10*.

FIGS. 6D and 6E show the effect of a locally maintained GPS-based timethat is late by 5 and 10 milliseconds, respectively, with respect to thesequence of expected data bits. Referring to FIG. 6D the start of the A1time segment (t=0 ms) is 5 milliseconds after the start of the exemplarysequence. Even though the local time has a time error of 5 milliseconds,all of the “A” time segments A1-11 and the “A” augmentation timesegments B4*, B8*, and B9* fall entirely within a data bit having thepolarity of the expected data bit for that time segment. The depolarizedincoming signal is integrated and when the correct code phase is used,the integration in the “A” time period is shown by a line referred to as“A” integration. The “A” integration increases monotonically during the“A” time segments A1-11 and “A” augmentation time segments B4*, B8*, andB9* to form the “A” time period magnitude. For example, during the A1time segment the “A” integration increases, during the B1 time segmentthe “A” integration is flat, during the A2 time segment the “A”integration increases, and so on until the “A” time period magnitude isreached. Note that the “A” integration increases during the “B” timesegments that are used as “A” augmentation time segments B4*, B8*, andB9*.

The B1 time segment in FIG. 6D starts 10 ms after (t=10 ms) after the A1time segment. The local time error of 5 milliseconds causes all of the“B” time segments B1-11 to straddle the transitions in the 20 ms databits. When the depolarized incoming signal is integrated and the correctcode phase is used, the integration in the “B” time period is shown by aline referred to as “B” integration. The “B” integration increasesduring the “B” augmentation time segments A5*, A9*, and A10 * but has nonet increase during the “A” and “B” time segments (except for the “A”time segments that are used as “B” augmentation time segments) to formthe “B” time period magnitude. For example, during the A1 time segmentthe “B” integration is flat, during the B1 time segment the “B”integration increases and then decreases for a net of zero, during theA2 time segment the “B” integration is flat, and so on until the “B”time period magnitude is reached. Note that the “B” integrationincreases during the “A” time segments that are used as “B” augmentationtime segments A5*, A9*, and A10 * and the “B” time segments that areused as “A” augmentation time segments B4*, B8*, and B9*.

Referring to FIG. 6E the start of the A1 time segment (t=0 ms) is 10milliseconds after the start of the exemplary sequence. Even though thelocal time has a time error of 10 milliseconds, all of the “A” timesegments A1-11 and the “A” augmentation time segments B4*, B8*, and B9*fall entirely within a data bit having the polarity of the expected databit for that time segment. The depolarized incoming signal is integratedand when the correct code phase is used, the integration in the “A” timeperiod is shown by a line referred to as “A” integration. The “A”integration increases monotonically during the “A” time segments A1-11and “A” augmentation time segments B4*, B8*, and B9* to form the “A”time period magnitude. For example, during the A1 time segment the “A”integration increases, during the B1 time segment the “A” integration isflat, during the A2 time segment the “A” integration increases, and soon until the “A” time period magnitude is reached. Note that the “A”integration increases during the “B” time segments that are used as “A”augmentation time segments B4*, B8*, and B9*.

The B1 time segment in FIG. 6E starts 10 ms after (t=10 ms) after the A1time segment. The local time error of 10 milliseconds causes all of the“B” time segments B1-11 to fall in the next 20 data bit. When thedepolarized incoming signal is integrated and the correct code phase isused, the integration in the “B” time period is shown by a line referredto as “B” integration. The “B” integration increases during the “B”augmentation time segments A5*, A9*, and A10 *, is flat during the “A”time segments (except for the “A” time segments that are used for “B”augmentation time segments) but decreases during the “B” time segments(except for the “B” time segments that are also used as “A” augmentationtime segments) to form the “B” time period magnitude. For example,during the A1 time segment the “B” integration is flat, during the B1time segment the “B” integration decreases, during the A2 time segmentthe “B” integration is flat, and so on until the “B” time periodmagnitude is reached. Note that the “B” integration increases during the“A” time segments that are used as “B” augmentation time segments A5*,A9*, and A10 * and the “B” time segments that are used as “A”augmentation time segments B4*, B8*, and B9*.

For the local GPS-based time that is late with respect to the correctGPS-based time as shown in FIGS. 6D and 6E the “A” time period magnitudeis greater than the “B” time period magnitude for 5 and 10 millisecondtime errors, respectively. For a local GPS time that is early withrespect to GPS-based time the time period magnitudes for “A” and “B”would be reversed and the “B” time period magnitude would be greaterthan the “A” time period magnitude.

FIG. 6F shows four types of time segments, termed “A”, “B”, “C”, and“D”, for an expected reception time period including the time period forthe expected sequence and early and late edge times before and after,respectively, the expected sequence. The “A” time segments are shown asA1, A3 , A5, A7 , A9, and A11; the “B” time segments are shown as B1, B3, B5 , B7 , B9, and B11; and so on for “C” and “D” time segments. Thefirst bit of the expected sequence is shown as bit 1; the third bit ofthe expected sequence is shown as bit 3; and so on with every other bitto bit 11. The time segments A1-11 odd, B1-11 odd, C1C1-11 odd, andD1-11 odd are shown in time with respect to the expected sequence for atime error of 0 ms. For a time error of 20 ms, the time segments A1-11odd, B1-11 odd, C1C1-11 odd, and D1-11 odd shift 20 ms right or leftwith respect to the expected sequence depending upon the sign of thetime error.

The A1, B1, C1, and D1 time segments are depolarized according to thesense of the known polarity for the expected bit 1; the A3 , B3 , C3,and D3 time segments are depolarized according to the sense of the knownpolarity for the expected bit 3; and so on to the A11 , B11 , C11, andD11 time segments depolarized according to the sense of the knownpolarity for the expected bit 11. The “A” time period magnitudes are thecoherent integrations corresponding to code phases, respectively, forthe A1, A3 , A5, A7 , A9, and A11time segments; the “B” time periodmagnitudes are the coherent integrations corresponding to code phases,respectively, for the B1, B3 , B5 , B7 , B9, and B11time segments; andso on for “C” and “D” time period magnitudes.

The four types of time segments (“A”, “B”, “C”, “D”) shown in FIG. 6Fcan be extended to N time segment types where the time segments of eachof the N time segment types alternates or follows one another as shownin FIG. 6F for N=4. Time periods for each of the N time segment typesare separately integrated. Any one of the N types of time segments canbe known as an Mth type of time segment. An Mth one of the time periodsincludes the integrations for the Mth type of time segments and mayinclude Mth augmentation time segments.

For N=8 having “A”, “B”, “C”, “D”, “E”, “F”, “G” and “H” time segments,there would be A1, A5, and A9 time segments; B1, B5 , and B9 timesegments and so on to H1, H5, and H9 time segments. For N=8,the A1through H1 time segments are arranged symmetrically about anddepolarized for the first bit (bit 1) in a sequence, the A5 through H5time segments are arranged symmetrically about and depolarized for thefifth bit (bit 5) in a sequence, and the A9 through H9 time segments arearranged symmetrically about and depolarized for the ninth bit (bit 9).

It should be seen by inspection that the time segments for at least oneof the “A”, “B”, “C”, and “D” time periods coincides with the expecteddata bits for up to a ±20 ms time error between the time segments basedupon the local estimate of GPS-based time and the sequence of expectedGPS data bits shown for actual GPS-based time. In addition, the sense ofthe known polarity of the data bits of the expected sequence does notchange for a time error of up to ±20 ms for the B9 and the C9 timesegments corresponding to the three consecutive data bits of bit 8, bit9, and bit 10. Therefore, the B9 and C9 time segments are noted with anasterisk (*) to show that they can be used as “A” augmentation timesegments and “D” augmentation time segments for determining the “A” and“D” time period magnitudes as described above for two types of timesegments (“A” and “B”). For N=2, “A” and “B” augmentation time segmentsresult from the “B” and “A” time segments corresponding to twoconsecutive data bits having the same sense; for N=4, “A” and “D”augmentation time segments result from the “B” and “C” time segmentscorresponding to three consecutive data bits having the same sense; forN=8 (“A”, “B”, “C”, “D”, “E”, “F”, “G” and “H”), “A”, “B”, “C”, “F”, “G”and “H” augmentation time segments result from the “D” and “E” timesegments corresponding five consecutive data bits having the same sense;and so on for N, N/2+1 consecutive data bits having the same senseresulting in augmentation time segments for the first N/2−1 and the lastN/2−1 types of time segments.

Doubling the number of time segment types halves the time accuracyrequirement. For example, the use of four types of time segments (“A”,“B”, “C”, “D”) instead of two types or time segment (“A”, “B”) reducesthe requirement for time accuracy from ±10 ms to ±20 ms; and the use ofeight types (N=8) reduces the requirement for time accuracy to ±40 ms.However, doubling the number of time segment types also reduces theprocessing gain by 3 dB. For example, the use of four time segment types(“A”, “B”, “C”, “D”) instead of two time segment types (“A”, “B”)reduces the processing gain by 3 dB.

FIGS. 7A-I are flow charts showing several arrangements for the order inwhich representations of the incoming GPS signal are depolarized,separated for “A” and “B” time segments, and integrated for providingthe “A” and “B” time period magnitudes for the potential code phases ofthe spreading code of the incoming signal. Although separations of “A”and “B” time segments are shown, it should be recognized that N types oftime segments can be separated, for example N=4 for “A”, “B”, “C”, and“D” time segment can be separated in the same way as for N=2 for “A” and“B” time segments. Several of the blocks in FIGS. 7A-I performintegrations of representations of the GPS signal. Those of ordinaryskill in the art will appreciate that such integrations can be performedas a single period of time or can be performed in separate periods oftime and then coherently accumulated.

In an embodiment shown in FIG. 7A, a block 702 depolarizes arepresentation of the incoming signal for the known polarities of theexpected GPS data bits for providing a depolarized signal. A block 704integrates the depolarized signal for time lengths not greater than thetime segments for providing depolarized integrations. A block 706separates the “A” time segments and the “B” time segments for thedepolarized integrated signal for providing “A” and “B” depolarizedintegrations, respectively. Then, a block 708 further integrates(accumulates) the “A” depolarized integrations over the entire “A” timeperiod for providing the “A” time period magnitudes and furtherintegrates (accumulates) the “B” depolarized integrations over theentire “B” time period for providing the “B” time period magnitudes.

In an embodiment shown in FIG. 7B, a block 710 depolarizes arepresentation of the incoming signal for the known polarities of theexpected GPS data bits for providing a depolarized signal. A block 712separates the “A” time segments and the “B” time segments for thedepolarized signal for providing an “A” depolarized signal and a “B”depolarized signal, respectively. Then, a block 714 integrates the “A”depolarized signal over entire “A” time period for providing the “A”time period magnitudes and integrates the “B” depolarized signal overthe entire “B” time period for providing the “B” time period magnitudes.

In an embodiment shown in FIG. 7C, a block 720 integrates arepresentation of the incoming signal for time lengths not greater thanthe time segments for providing first integrations of the signal. Ablock 722 depolarizes the first integrations for the known polarities ofthe expected GPS data bits for providing depolarized integrations. Ablock 724 separates the depolarized integrations according to the “A”and “B” time segments for providing “A” and “B” depolarizedintegrations, respectively. Then, a block 726 further integrates the “A”depolarized integrations over the entire “A” time period for providingthe “A” time period magnitudes and further integrates the “B”depolarized integrations over the entire “B” time period for providingthe “B” time period magnitudes.

In an embodiment shown FIG. 7D, a block 730 integrates a representationof the incoming signal for time lengths not greater than the timesegments for providing first integrations of the signal. A block 732separates the first integrations according to the “A” and “B” timesegments for providing “A” and “B” depolarized integrations,respectively. A block 734 depolarizes the “A” and “B” integrations forthe known polarities of the expected GPS data bits for providing “A” and“B” depolarized integrations, respectively. Then, a block 736 furtherintegrates the “A” depolarized integrations over the entire “A” timeperiod for providing the “A” time period magnitudes and furtherintegrates the “B” depolarized integrations over the entire “B” timeperiod for providing the “B” time period magnitudes.

In an embodiment shown in FIG. 7E, a block 740 separates arepresentation of the incoming signal according to “A” and “B” timesegments for providing “A” and “B” time segment signals, respectively. Ablock 742 depolarizes the “A” and “B” time segment signals for the knownpolarities of the expected GPS data bits for providing “A” and “B”depolarized time segment signals, respectively. Then, a block 744integrates the “A” depolarized time segment signal over the entire “A”time period for providing the “A” time period magnitudes and integratesthe “B” depolarized time segment signal over the entire “B” time periodfor providing the “B” time period magnitudes.

An embodiment shown in FIG. 7F, a block 750 separates a representationof the incoming signal according to “A” and “B” time segments forproviding “A” and “B” time segment signals, respectively. A block 752integrates the “A” and “B” time segment signals for time lengths notgreater then time segments for providing “A” and “B” integrations,respectively. A block 754 depolarizes the “A” and “B” integrations forthe known polarities of the expected GPS data bits for providing “A” and“B” depolarized integrations, respectively. Then, a block 756 furtherintegrates the “A” depolarized integrations over the entire “A” timeperiod for providing the “A” time period magnitudes and furtherintegrates the “B” depolarized integrations over the entire “B” timeperiod for providing the “B” time period magnitudes.

In an embodiment shown in FIG. 7G, a block 760 integrates arepresentation of the incoming signal for time lengths less than thetime segments for providing first integrations, for example one epoch (1ms for GPS C/A code). A block 762 depolarizes the first integrations forthe known polarities of the expected GPS data bits for providingdepolarized first integrations. A block 764 further integrates thedepolarized first integrations for time lengths not greater than thetime segments, preferably equal to a time segment (10 ms for GPS C/Acode), for providing depolarized second integrations of the signal. Ablock 766 separates the depolarized second integrations according to “A”and “B” time segments for providing “A” and “B” depolarized secondintegrations, respectively. Then, a block 768 further integrates the “A”depolarized second integrations over the entire “A” time period forproviding the “A” time period magnitudes and further integrates the “B”depolarized second integrations over the entire “B” time period forproviding the “B” time period magnitudes. The embodiment of FIG. 7Gdiffers from the embodiments of FIGS. 7A, 7C, 7D, BE, and 7I in that arepresentation of the GPS signal is depolarized between integrationblocks for performing time segment integrations. Those of ordinary skillin the art will note that the embodiment of 7G effectively reduces toone of the embodiments illustrated in FIGS. 7A, 7C, 7D, BE, and 7I whenthe depolarization block 762 is placed elsewhere than between theintegrations blocks 760 and 764. For example, when the block 762 ismoved to a position before the block 760, the embodiment of FIG. 7Greduces to the embodiment illustrated in FIG. 7A.

In an embodiment shown in FIG. 7H, a block 770 integrates arepresentation of the incoming signal for time lengths less than thetime segments for providing first integrations, for example one epoch (1ms for GPS C/A code). A block 772 separates the first integrationsaccording to “A” and “B” time segments for providing “A” and “B” firstintegrations, respectively. A block 774 depolarizes the “A” and “B”first integrations for the known polarities of the expected GPS databits for providing “A” and “B” depolarized integrations, respectively.Then, a block 776 further integrates the “A” depolarized integrationsover the entire “A” time period for providing the “A” time periodmagnitudes and further integrates the “B” depolarized integrations overthe entire “B” time period for providing the “B” time period magnitudes.

In an embodiment shown in FIG. 7I, a block 780 integrates arepresentation of the incoming signal for time lengths less than thetime segments for providing first integrations, for example one epoch (1ms for GPS C/A code). A block 782 depolarizes the first integrations forproviding depolarized integrations. A block 784 separates thedepolarized integrations according to “A” and “B” time segments forproviding “A” and “B” depolarized integrations, respectively. Then, ablock 786 further integrates the “A” depolarized integrations over theentire “A” time period for providing the “A” time period magnitudes andfurther integrates the “B” depolarized integrations over the entire “B”time period for providing the “B” time period magnitudes.

FIG. 8 is a block diagram of an N processor referred to by a referencenumber 800 for processing N types of time segments. The N processor 800for a special case of N=2 is described above as the AB processor 136.FIG. 8 illustrates the N processor 800 for N=4 for processing “A”, “B”,“C”, and “D” types of time segments. The “A”, “B”, “C”, and “D” types oftime segments are illustrated in FIG. 6F and described in theaccompanying detailed description. The N processor 800 receives I and Qintegrations from the I and Q accumulators 166 of the time domaincorrelation machine 112A or the IFFT 180 of the frequency domaincorrelation machine 112B. One of the depolarizers 143-149 or 172 is inplace for depolarizing the representation of the incoming signal each ofthe N types of time segments.

The N processor 800 includes an N switch 802, I accumulators 804, Qaccumulators 805, and combiners 806. The N switch 802 separates the Iand Q integrations for each of the N types of time segments into N typesof time periods denoted for N=4 as “A”, “B”, “C”, and “D” time periods.The “A” time period includes “A” time segments and “A” augmentation timesegments (FIG. 6F); the “B” time period includes “B” time segments and“B” augmentation time segments; and so on for the “C” and “D” timesegments and augmentation time segments. There are several variations(FIGS. 7A-I) regarding the placement of the N switch 802 within the GPSreceiver 100 for separating the incoming signal. In one variation the Nswitch 802 separates the stored signal representation where the storedsignal representation enters the correlation machine 112 from the signalmemory 108. In this variation the correlation machine 112 integrates thestored signal representation for each of the N types of time segmentsand augmentation time segments separately from each of the other of theN types of time segments and augmentation time segments. In anothervariation the N switch 802 is disposed within the correlation machine112 so that signal can be processed in time periods of up to the 10 mstime periods of the time segments before the N switch 802.

In an embodiment, the N switch 802 includes an I N switch 802I and a Q Nswitch 802Q. In FIG. 8, the I N switch 802I and the Q N switch 802Q areshown for the “B” time period. The I N switch 802I receives thedepolarized I integrations from the I accumulator 166 for the timedomain correlation machine 112A or the IFFT 180 for the frequency domaincorrelation machine 112B. Similarly, the Q N switch 802Q receives thedepolarized Q integrations from the Q accumulator 166 for the timedomain correlation machine 112A or the IFFT 180 for the frequency domaincorrelation machine 112B.

For N=4, the I accumulators 804 include four I accumulators denoted asan “A” I accumulator 804A, a “B” I accumulator 804B, a “C” I accumulator804C, and a “D” I accumulator 804D. Similarly, the Q accumulators 805include four Q accumulators denoted as an “A” Q accumulator 805A, a “B”Q accumulator 805B, a “C” Q accumulator 805C, and a “D” Q accumulator805D. For the “A” time period, the I N switch 802I passes thedepolarized I integrations to the “A” I accumulator 804A; for the “B”time period, the I N switch 802I passes the depolarized I integrationsto the “B” I accumulator 804B; for the “C” time period, the I N switch802I passes the depolarized I integrations to the “C” I accumulator804C; and for the “D” time period, the I N switch 802I passes thedepolarized I integrations to the “D” I accumulator 804D. Similarly, forthe “A” time period, the Q N switch 802Q passes the depolarized Qintegrations to the “A” Q accumulator 805A; for the “B” time period, theQ N switch 802Q passes the depolarized Q integrations to the “B” Qaccumulator 805B; for the “C” time period, the Q N switch 802Q passesthe depolarized Q integrations to the “C” Q accumulator 805C; and forthe “D” time period, the Q N switch 802Q passes the depolarized Qintegrations to the “D” Q accumulator 805D. The accumulators 804A-Daccumulate the depolarized I integrations for providing I magnitudeaccumulations for each code phase. Similarly, the accumulators 805A-Daccumulate the depolarized Q integrations for providing Q magnitudeaccumulation for each code phase.

For N=4, the N combiner 806 includes an “A” combiner 806A, a “B”combiner 806B, a C combiner 806C, and a “D” combiner 806D. The “A”combiner 806A adds the square of the “A” time period I accumulation fromthe “A” I accumulator 804A to the square of the “A” time period Qaccumulation from the “A” Q accumulator 805A for each code phaseincrement for providing the “A” time period magnitudes; the “B” combiner806B adds the square of the “B” time period I accumulation from the “B”I accumulator 804B to the square of the “B” time period Q accumulationfrom the “B” Q accumulator 805B; and so on for the “C” combiner 806C andthe “D” combiner 806D. At this point using code phase increments ofslightly less than one-half chip there will be preferably about 2048 “A”time period magnitudes, about 2048 “B” time period magnitudes, about2048 “C” time period magnitudes, and about 2048 “D” time periodmagnitudes. As described above, the code phase increment that results inthe strongest of the “A”, “B”, “C”, or “D” time period magnitudes, orcombination of the “A”, “B”, “C”, and “D” time period magnitudes for acode phase is tested against a correlation threshold and used for signalacquisition when the correlation threshold is exceeded.

As described above the time period I and Q accumulations are preferablyproportional to the levels for the I and Q integrations of the GPSsignal, whereas the “A”, “B”, “C”, and “D” time period magnitudes arepreferably proportional to the sum of the squares the I and Q magnitudeaccumulations (I²+Q²). However, it should be noted that the “A”, “B”,“C”, and “D” time period magnitudes can be some other convenientnon-decreasing functions of the time period I and Q accumulations, suchas the sum (I+Q) or the square root of the sum of the squares(√(I²+Q²)).

FIG. 9 is a block diagram of a signal receiver referred to by thegeneral reference number 810. The receiver 810 is described in terms ofa global positioning system (GPS) receiver for receiving incomingcoarse/acquisition (C/A) GPS signals. However, those of ordinary skillthe art of spread spectrum radio receivers will see that the blockdiagram illustrated in FIG. 9 and described below can be applied forreceiving other types of signals, especially for receiving other typesof direct fast sequence spread spectrum signals including precision P orP(Y) code GPS signals. The GPS receiver 810 includes the GPS antenna102, the frequency downconverter 104, the reference generator 105, thesampler 106, the stored signal memory 108, the correlation machine 112,the microprocessor 114, the replica generator 138, and the programmemory 116 including the stored sequence 122, the acquisition detector811, and the signal and navigation program 126 as described above.

The GPS receiver 810 also includes an expected sequence timer 812, andan N timer 814. The sequence timer 812 uses the reference generator 105for maintaining GPS time to within ±20/N ms for gating the sampler 106and/or the signal memory 108 for storing the sampled downconverted GPSsignal during the expected reception time period including the early andlate edge times as shown in FIG. 6F. The sequence timer 812 alsotriggers the N timer 814 for generating N time segment signals fororganizing the expected reception time period into N types ofalternating 10 ms time segments as illustrated in FIGS. 6A-E for N=2 andFIG. 6F for N=4 and described in the accompanying detailed descriptionsabove.

The correlation machine 112 includes the N processor 800 illustrated inFIG. 8 and described above for the case of N=4. The correlation machine112 including the N processor 800 integrates the stored signalrepresentation of the GPS signal during the “A” time segments and “A”augmentation time segments for providing “A” time period magnitudes foreach code phase, respectively; during the “B” time segments and “B”augmentation time segments for providing “B” time period magnitudes foreach code phase, respectively; during the “C” time segments and “C”augmentation time segments for providing “C” time period magnitudes foreach code phase, respectively; and during the “D” time segments and “B”augmentation time segments for providing for “D” time period magnitudeseach code phase, respectively. The correlation machine 112 andintegration process are illustrated and described in greater detail inFIGS. 3-4 and the accompanying descriptions above with the provisionthat the AB processor 136 is replaced by the N processor 800.

The N types of time period magnitudes, represented for N=4 by the “A”time period magnitudes, the “B” time period magnitudes, “C” time periodmagnitudes and the “D” time period magnitudes, from the correlationmachine 112 are processed by the microprocessor 114 using instructionsin the acquisition detector 811 for determining the increment of codephase for the replica code that appears to match the code phase of theincoming GPS signal. This replica code phase is then processed for GPSsignal acquisition by the microprocessor 114 according to instructionsin the signal and navigation program 126.

There are several embodiments for the acquisition detector 811 that canprocess the N types of time period magnitudes in order to find the codephase for acquisition. In an embodiment, the strongest of the timeperiod magnitudes is tested against a correlation threshold. In anotherembodiment, the N types of time period magnitudes for a code phase arecombined (incoherently) or their squares are added to form an N combinedtime period magnitude for that code phase. The code phase that resultedin the strongest of the time period magnitudes is tested against acorrelation threshold. When the strongest time period magnitude exceedsthe correlation threshold, the code phase that resulted in that combinedtime period magnitude is used for signal acquisition and tracking. Oncethe code phase for acquisition of the GPS signal from a single GPSsatellite is determined, conventional methods can be used for acquiringthe GPS signal from other GPS satellites. When the strongest magnitudedoes not exceed the required correlation threshold, then a differentDoppler frequency, a different PRN code, a different GPS time, or thelike is tried and the process is repeated; or the GPS receiver 810 maygo into a standby mode to conserve power.

FIG. 10 is a block diagram of a signal receiver referred to with ageneral reference number 1000. The receiver 1000 is described in termsof a global positioning system (GPS) receiver for receiving incomingcoarse/acquisition (C/A) GPS signals. However, those of ordinary skillin the art of spread spectrum radio receivers will see that the blockdiagram illustrated in FIG. 10 and described below can be applied forreceiving direct sequence spread spectrum (DSSS) signals in general andespecially for receiving GPS signals of other codes such the P or P/Ycode, and for receiving Galileo signals, GLONASS signals, globalnavigation system satellite (GNSS) signals in general, and communicationsystem CDMA signals.

The GPS receiver 1000 includes the GPS antenna 102, the frequencydownconverter 104 including the reference generator 105, the sampler106, the optional signal memory 108, the correlation machine 112including the AB processor 136 and the replica generator 138, themicroprocessor 114, the hardware for the program memory 116 having thefirmware for the acquisition detector 124 and the signal and navigationprogram (SNP) 126, and the AB timer 134 that have been described above.The receiver 1000 also includes a data bit search generator 1002 forgenerating trial-and-error data bit search patterns that are used in thecorrelation machine 112. The correlation machine 112 uses the data bitsearch patterns for depolarizing the data bits of the incoming signalover several data bit time periods for determining an “A” magnitude setand a “B” magnitude set for each search pattern. The “A” magnitude setis a set of “A” magnitudes corresponding to code phase increments,respectively, of the incoming spread spectrum signal. Similarly, the “B”magnitude set is a set of “B” magnitudes for code phase increments,respectively, of the incoming spread spectrum signal. The data bit testpatterns are illustrated described in FIGS. 12A and 12B for two andthree data bit sequences and described in the accompanying detaileddescriptions.

FIG. 11A is a flow chart of a method for acquiring spread spectrumsignals using a trial-and-error data bit search with “A” and “B” timeperiods. The steps in the method may be embedded in a computer-readableform on a tangible medium 1100 that may be read by a computer, such asthe microprocessor 114 in the GPS receiver 1000, for carrying out thesteps. In a step 52 the incoming GPS signal is received. In a step 54the GPS signal is downconverted and sampled.

Data bit search patterns are generated in a step 1102 for atrial-and-error search for determining a data bit pattern that matchesthe data bits carried in the incoming signal. The number of data bitsearch patterns is 2^((n−1)) where n is number of data bit time lengthsin the reception time length. In a step 1104 a reception time length forthe incoming signal is established and organized into alternating “A”time segments and “B” time segments.

A representation of the downconverted, sampled incoming signal isdepolarized in a step 1106 with the data bit search patterns to providedata bit search result signals. The incoming signal representation maybe the downconverted sampled incoming signal directly as it is receivedin real-time from the step 54, or the representation may be thedownconverted sampled incoming signal after it is stored in the signalmemory 108. When the signal is stored in the signal memory 108 the samestored signal may be used to test for a first data bit search patternand then retested for a second, third, fourth, and so on, data bitsearch pattern. The FIGS. 7A-I shows the blocks 702, 710, 722, 734, 742,754, 762, 774 or 782 for the signal flow position where the incomingsignal representation is depolarized. FIG. 13 shows structural blocks143-149 for depolarizing the signal representation for a time domainsystem. FIG. 14 shows structural blocks 143, 144, 146, 147 or 172 fordepolarizing the signal representation for a frequency domain system.

The search result signals for the “A” time segments are integrated in astep 1112 for determining “A” magnitude sets. Each data bit searchpattern results in a set of “A” magnitudes (also called correlationmagnitudes or time period magnitudes herein). Each “A” magnitudecorresponds to the correlation between a replica signal (from thereplica generator 138) and the downconverted sampled signal (from thesampler 106) for a code phase increment for the replica signal.Similarly, the search result signals for the “B” time segments areintegrated in a step 1114 for determining “B” magnitude sets. Each databit search pattern results in a set of “B” magnitudes (also calledcorrelation magnitudes or time period magnitudes herein). Each “B”magnitude corresponds to the correlation between a replica signal (fromthe replica generator 138) and the downconverted sampled signal (fromthe sampler 106) for a code phase increment for the replica signal.FIGS. 7A-I shows blocks 708, 714, 726, 736, 746, 756, 768, 778 or 786for the signal flow for integrating the data bit search result signals.

It should be noted that for incoming signal acquisition, it may benecessary to repeat the process of steps 1106, 1112 and 1114 for eachpotential carrier frequency.

For example, a first data bit search pattern is generate in the step1102. The first pattern is used to depolarize the incoming signalrepresentation in the step 1106 to provide a first data bit searchresult signal. In the step 1112 correlations are determined on the firstdata bit search result for each code phase increment and thecorrelations are accumulated for the code phase increment throughout the“A” time segments in the reception time length to provide the “A”magnitudes for the code phase increments, respectively. Similarly, inthe step 1114 correlations are determined on the first data bit searchresult for each code phase increment and the correlations areaccumulated for the code phase increment throughout the “B” timesegments in the reception time length to provide the “B” magnitudes forthe code phase increments, respectively.

Then, a second data bit search pattern is generated in the step 1102.The second pattern is used to depolarize the incoming signalrepresentation in the step 1106 to provide a second data bit searchresult signal. In the step 1112 correlations are determined on thesecond data bit search result for each code phase increment and thecorrelations are accumulated for the code phase increment throughout the“A” time segments in the reception time length to provide the “A”magnitudes for the code phase increments, respectively. Similarly, inthe step 1114 correlations are determined on the second data bit searchresult for each code phase increment and the correlations areaccumulated for the code phase increment throughout the “B” timesegments in the reception time length to provide the “B” magnitudes forthe code phase increments, respectively. This process is continued forhowever many patterns are required. The result is an “A” magnitude foreach data bit pattern for each code phase increment and a “B” magnitudefor each data bit pattern for each code phase increment.

The process of accumulating correlations is continued for each data bitsearch pattern until “A” and “B” magnitude sets have been determined forall 2^((n−1)) search patterns where n is the number of data bit timelengths in the signal reception time length.

The code phase increment that results in the strongest of the magnitudes(the greatest correlation) is identified in the step 70. Then, in a step1116 the code phase increment corresponding to the strongest magnitudeis used for receiving the incoming signal so that the signal can beacquired and/or tracked after the signal is acquired in order todetermine the location where the signal is received.

The “A” depolarized signal representations are integrated (coherentlyaccumulated) in the step 1112 for an “A” time period including all the“A” time segments for providing the “A” (time period) magnitudes.Similarly, in the step 1114 the “B” depolarized signal representationsare integrated (coherently accumulated) for a “B” time period includingall the “A” time segments for providing the “B” (time period)magnitudes. The signal integrations for the “A” time period provide “A”magnitudes corresponding to the potential code phase increments,respectively, and the signal integrations for the “B” time periodprovide “B” magnitudes corresponding to the same potential code phaseincrements, respectively. Then, in the step 70 described above, the codephase increment that results in the strongest of the “A” or “B”magnitudes is tested against a correlation threshold and used for signalacquisition and/or tracking when a correlation threshold is exceeded. Inorder to find a time period magnitude that exceeds the correlationthreshold the steps may need to be iterated using different assumptionsfor carrier frequency.

FIG. 11B is a flow chart of a method for acquiring spread spectrumsignals using a trial-and-error data bit search with “A” and “B” timeperiods. The steps in the method may be embedded in a computer-readableform on a tangible medium 1150 that may be read by a computer, such asthe microprocessor 114 in the GPS receiver 1000, for carrying out thesteps. The operation of an embodiment can be stated as follows.

Step 1152. Organize and establish a reception time length of two or moredata bits into a series of alternating A and B time segments, each timesegment being 1/2 data bit time (in C/A code the time segments are 10 msand the time length of 2 or 3 data bits is 40 or 60 ms). Either A timesegments or B time segments will always avoid data bit transitions.

Step 1154. Organize and establish an A non-continuous time period toinclude all A time segments and a B non-continuous time period toinclude all B time segments. The “A” and “B” time segments alternate.

Step 1156. Perform coherent integration for each code phase increment(with each carrier frequency if necessary) for each data bit sequencecase for the A time period and the B time period to accumulate themagnitude of the correlation level for each code phase increment (foreach carrier frequency if necessary) for each data bit sequence case(each data bit search pattern) for the A time period and a correlationlevel for each code phase increment (for each carrier frequency ifnecessary) for each data bit sequence case for the B time period. Thestep 1156 can be performed with standard time domain search or frequencytransforms or combination of time domain search and frequencytransforms. The coherent integration is performed by correlatingspreading code chips of the incoming signal representation withspreading code chips of a replica signal at each code phase incrementand accumulating the correlations throughout the “A” time period andindependently throughout the “B” time period.

Step 1158. Use the correlation levels to select the correct code phaseincrement. For example, choose the code phase increment (and carrierfrequency) from the sequence (data bit search pattern) from either the Aor B time period that corresponds to the strongest—largest absolutevalue—correlation level.

FIG. 12A is a time chart showing “A” and “B” time segments and showingthe accumulated “A” and “B” magnitudes for a reception time lengthhaving two data bit time lengths. For the GPS C/A the data bit timelength is 20 milliseconds. The “A” and “B” time segment lengths areone-half of the data bit time length. For the GPS C/A code, the timesegment length is 10 milliseconds. An exemplary actual, but unknown,data bit pattern for the incoming signal is shown as 01 followed by a 0.The timing is arbitrarily shown for the data bit transitions fallingmidway during the “B” time segments.

The data bit generator 1002 generates two data bit search patterns—a 1stdata bit pattern of 00 or its inverse of 11, and a 2nd data bit patternof 01 or its inverse of 10. Because either a strong positive or a strongnegative value for magnitude equally shows correlation, it is notnecessary to generate or test both a pattern and its the inversepattern. Each of the two patterns—00 and 01—are used for depolarizingthe incoming signal representation for determining data bit searchresults.

The accumulated “A” and “B” magnitudes are shown for the correct codephase increment (and carrier frequency) when the data bit search patternis correct (the 2nd data bit search pattern in this example). The “A”magnitude accumulates to an increasingly greater value as thecorrelations for more code epochs (for GPS C/A there are 20 code epochsper data bit) are accumulated for the correct data bit search pattern atthe correct code phase increment (and carrier frequency). The “B”magnitudes, even for the correct data bit search pattern and the correctcode phase increment (and carrier frequency) do not accumulate astronger value when a data bit transition inverts the accumulationbecause the portion of the accumulation after the transition subtractsfrom the portion of the accumulation before the transition.

Case 00 or 11. Assume no transition. Perform coherent integrationthroughout first A time segment, do not invert, then continue coherentintegration throughout second A time segment to determine correlationlevels for the A time period. Similarly for B time segments.Case 01 or 10. Assume a transition. Perform coherent integrationthroughout first A time segment, invert, then continue coherentintegration throughout second A time segment to determine correlationlevels for A time period. Similarly for B time segments.

FIG. 12B is a time chart showing “A” and “B” time segments and showing“A” and “B” magnitudes for a reception time length having three data bittime lengths. For the GPS C/A the data bit time length is 20milliseconds. The “A” and “B” time segment lengths are one-half of thedata bit time length. For the GPS C/A the time segment length is 10milliseconds. An exemplary actual, but unknown, data bit pattern for theincoming signal is shown as 101 followed by a 0. The timing isarbitrarily shown for the data bit transitions falling midway during the“B” time segments.

The data bit generator 1002 generates four data bit search patterns—a1st data bit pattern of 000 or its inverse 111, a 2nd data bit patternof 001 or its inverse 110, a 3rd data bit pattern of 101 or its inverse010, and a 4th data bit pattern of 011 or its inverse 100. Becauseeither a strong positive or a strong negative value for magnitudeequally shows correlation, it is not necessary to generate or test botha pattern and its the inverse. Each of the four patterns—000, 001, 101and 011—are used for depolarizing the incoming signal representation fordetermining data bit search results.

The “A” and “B” magnitudes are shown for the correct code phaseincrement (and carrier frequency) when the data bit search pattern iscorrect (the 3rd data bit search pattern in this example). The “A”magnitude accumulates to an increasingly greater value as thecorrelations for more code epochs (for GPS C/A there are 20 code epochsper data bit) are accumulated for the correct data bit search pattern atthe correct code phase increment (and carrier frequency). The “B”magnitudes, even for the correct data bit search pattern and the correctcode phase increment (and carrier frequency) do not accumulate astronger value when a data bit transition inverts the accumulation sothat the portion of the accumulation after the transition subtracts fromthe portion of the accumulation before the transition.

Case 000 or 111) Perform coherent integration throughout first A timesegment; do not invert, then continue coherent integration throughoutsecond A time segment; do not invert, then continue coherent integrationthroughout third A time segment to determine correlation levels for Atime period. Similarly for B time segments to determine correlationlevels for B time period.Case 001 or 110) Perform coherent integration throughout first A timesegment; do not invert, then continue coherent integration throughoutsecond A time segment; invert, then continue coherent integrationthroughout third A time segment to determine correlation levels for Atime period. Similarly for B time segments to determine correlationlevels for B time period.Case 010 or 101) Perform coherent integration throughout first A timesegment; invert, then continue coherent integration throughout second Atime segment; invert, then continue coherent integration throughoutthird A time segment to determine correlation levels for A time period.Similarly for B time segments to determine correlation levels for B timeperiod.Case 011 or 100) Perform coherent integration throughout first A timesegment; invert, then continue coherent integration throughout second Atime segment; do not invert, then continue coherent integrationthroughout third A time segment to determine correlation level levelsfor A time period. Similarly for B time segments to determinecorrelation levels for B time period.

FIG. 13 is a block diagram showing the signal memory 108, the timedomain correlation machine 112A (FIG. 3) and the data bit searchgenerator 1002. The structures, functions and results of the elements ofthe correlation machine 112A are described above in detail. The data bitgenerator 1002 generates the data bit search patterns used by one of theelements 143, 144, 145, 146, 147, 148 or 149 for depolarizing theincoming signal representation.

FIG. 14 is a block diagram showing the signal memory 108, the frequencydomain correlation machine 112B (FIG. 4) and the data bit searchgenerator 1002. The structures, functions and results of the correlationmachine 112B are described above in detail. The data bit generator 1002generates the data bit search patterns used by one of the elements 143,144, 146, 147 or 172 for depolarizing the incoming signalrepresentation.

FIG. 15 is a flow chart for an embodiment using the GPS C/A code as anexemplary case. The steps in the method may be embedded in acomputer-readable form on a tangible medium 1500 that may be read by acomputer, such as the microprocessor 114 in the receiver 1000, forcarrying out the steps. The method may be implemented in the correlationmachines 112A or 112B. In a step 1502 the code correlations for offsetsof each code phase increment between the spreading codes of a replicasignal and the incoming signal are accumulated for I and Q for one-halfthe time length of a data bit for a first “A” time segment. The first“A” time segment accumulations for I and Q are remembered for each codephase increment. Similarly, in a step 1506 the code correlations foroffsets of each code phase increment between the spreading codes of areplica signal and the incoming signal are accumulated for I and Q forone-half the time length of a data bit for a first “B” time segment. Thefirst “B” time segment accumulations are remembered for I and Q for eachcode phase increment.

In a step 1512 the code correlations for offsets of each code phaseincrement between the spreading codes of a replica signal and theincoming signal are accumulated for I and Q for one-half the time lengthof a data bit for a second “A” time segment. The second “A” time segmentaccumulations for I and Q are remembered for each code phase increment.Similarly, in a step 1516 the code correlations for offsets of each codephase increment between the spreading codes of a replica signal and theincoming signal are accumulated for I and Q for one-half the time lengthof a data bit for a first “B” time segment. The second “B” time segmentaccumulations are remembered for I and Q for each code phase increment.

In a step 1522 the second “A” time segment accumulations are depolarizedaccording to the first data bit search pattern. The depolarizationinverts the second “A” time segment accumulations when the second bit ofthe first search pattern is different than the first bit. When thesecond bit of the first search pattern is the same as the first bit, thedepolarization does not invert the second “A” time segmentaccumulations. For example the data bit search pattern of 01 would causethe second “A” time segment accumulations to be inverted and the databit search pattern of 00 would not cause second “A” time segmentaccumulations to be inverted. Similarly, the second “B” time segmentaccumulations are depolarized according to the first data bit searchpattern. The depolarization inverts the second “B” time segmentaccumulations when the second bit of the first search pattern isdifferent than the first bit. When the second bit of the first searchpattern is the same as the first bit, the depolarization does not invertthe second “B” time segment accumulations.

In a step 1524, for the first data bit search pattern, the I first “A”time segment accumulations are added to the depolarized second “A” timesegment accumulations for I for each code phase increment, respectively.The Q first “A” time segment accumulations are added to the Qdepolarized second “A” time segment accumulations for each code phaseincrement, respectively. Similarly, the I first “B” time segmentaccumulations are added to the I depolarized second “B” time segmentaccumulations for each code phase increment, respectively. The Q first“B” time segment accumulations are added to the Q depolarized second “B”time segment accumulations for each code phase increment, respectively.In a step 1526, for the first search pattern, first “A” magnitudes arecomputed from the sum of the squares of the I and Q for each code phaseincrement. Similarly, for the first search pattern, first “B” magnitudesare computed from the sum of the squares of the I and Q for each codephase increment.

In a step 1532 the second “A” time segment accumulations are depolarizedaccording to the second data bit search pattern. The depolarizationinverts the second “A” time segment accumulations when the second bit ofthe second search pattern is different than the first bit. When thesecond bit of the second search pattern is the same as the first bit,the depolarization does not invert the second “A” time segmentaccumulations. Similarly, the second “B” time segment accumulations aredepolarized according to the second data bit search pattern. Thedepolarization inverts the second “B” time segment accumulations whenthe second bit of the second search pattern is different than the firstbit. When the second bit of the second search pattern is the same as thefirst bit, the depolarization does not invert the second “B” timesegment accumulations.

In a step 1534, for the second data bit search pattern, the I first “A”time segment accumulations are added to the I depolarized second “A”time segment accumulations for each code phase increment, respectively.The Q first “A” time segment accumulations are added to the Qdepolarized second “A” time segment accumulations for each code phaseincrement, respectively. Similarly, the I first “B” time segmentaccumulations are added to the I depolarized second “B” time segmentaccumulations for each code phase increment, respectively. The Q first“B” time segment accumulations are added to the Q depolarized second “B”time segment accumulations for each code phase increment, respectively.In a step 1536, for the second search pattern, second “A” magnitudes arecomputed from the sum of the squares of the I and Q for each code phaseincrement. Similarly, for the second search pattern, second “B”magnitudes are computed from the sum of the squares of the I and Q foreach code phase increment.

When the data bit search pattern time length is n data bits where n isgreater than two, the above steps are repeated in a step 1540 foraccumulating the I and Q code correlations at the code phase incrementsfor the code epochs in the “A” and “B” time segments for each of thedata bits up to n; depolarizing the accumulations according to thesenses of the nth data bit of the search pattern; adding the depolarizedaccumulations for the 2^(n−1) search patterns, respectively; andcomputing the 1 to 2^(n−1) “A” and “B” magnitudes from the sums of thesquares of the I and Q of the added depolarized accumulations.

In a step 1542 the strongest (generally the largest) of the firstthrough 2^(n−1) “A” or “B” magnitudes is determined. The particular codephase increment that corresponds to the largest of these magnitudes isused for receiving the incoming signal, acquiring the incoming signal,tracking of the incoming signal, and/or determining the time-of-arrivalof the incoming signal in order to determine the location where thesignal is being received.

Although the present invention has been described in terms of thepresently preferred embodiments, it is to be understood that suchdisclosure is not to be interpreted as limiting. Various alterations andmodifications will no doubt become apparent to those skilled in the artafter having read the above disclosure. Accordingly, it is intended thatthe broadest reading of the appended claims define the true idea andscope of the invention.

The below listed claims derive support from and must be read in view ofthe whole specification and all the drawing figures.

1. A method implemented in a receiver for receiving an incoming signalhaving data bits spread by a spreading code, comprising: integratingsaid incoming signal at code phase increments of said spreading codewith two or more data bit search patterns in an “A” integration timeperiod comprising at least two non-contiguous “A” time segments fordetermining magnitudes corresponding to said code phase increments,respectively, for each of said search patterns; integrating saidincoming signal at said code phase increments with said search patternsfor a “B” integration time period comprising “B” time segmentsalternating with said “A” time segments for determining magnitudescorresponding to said code phase increments, respectively, for each ofsaid search patterns; and using a largest of said magnitudes fordetermining a particular one of said code phase increments for receivingsaid incoming signal.
 2. The method of claim 1, further comprising:generating 2^((n−1)) said search patterns where said n is the number ofdata bit time lengths in a signal reception time length.
 3. The methodof claim 1, further comprising: establishing said alternating “A” and“B” time segments for a signal reception time length of two or more saiddata bits; and wherein: integrating includes depolarizing arepresentation of said incoming signal with said search patterns forproviding “A” and “B” said magnitudes corresponding to said “A” and “B”integration time periods, respectively.
 4. The method of claim 1,wherein: said search patterns include 00 or its inverse and 01 or itsinverse when a signal reception time length is about two said data bittime lengths.
 5. The method of claim 1, wherein: said search patternsinclude 000 or its inverse, 001 or its inverse, 010 or its inverse and100 or its inverse when a signal reception time length is about threedata bit time lengths.
 6. The method of claim 1 wherein: a first of saidsearch patterns has a no inversion between a first and a second bit forproviding first “A” said magnitudes for said “A” integration time periodand first “B” said magnitudes for said “B” integration time period; anda second of said search patterns has an inversion between said first andsaid second bit for providing second “A” said magnitudes for said “A”integration time period and second “B” said magnitudes for said “B”integration time period.
 7. The method of claim 1, wherein: a first ofsaid search patterns has no inversion between a first and a second bitand no inversion between said second and a third bit for providing first“A” said magnitudes for said “A” integration time period and first “B”said magnitudes for said “B” integration time period; a second of saidsearch patterns has no inversion between said first and said second bitand an inversion between said second and said third bit for providingsecond “A” said magnitudes for said “A” integration time period andsecond “B” said magnitudes for said “B” integration time period; a thirdof said search patterns has an inversion between said first and saidsecond bit and an inversion between said second and said third bit forproviding third “A” said magnitudes for said “A” integration time periodand third “B” said magnitudes for said “B” integration time period; anda fourth of said search patterns has an inversion between said first andsaid second bit and no inversion between said second and said third bitfor providing fourth “A” said magnitudes for said “A” integration timeperiod and fourth “B” said magnitudes for said “B” integration timeperiod.
 8. The method of claim 1, wherein: said time segments have timelengths about one-half a time length of said data bits.
 9. The method ofclaim 1, wherein: said incoming signal is a global navigation satellitesystem (GNSS) signal.
 10. The method of claim 1; wherein: integratingincludes performing code phase increment-by-increment code correlationsbetween and said spreading code of said incoming signal and a localreplica of said spreading code; and accumulating said code correlationsthrough several code epochs of said spreading code during said “A” timesegments for determining “A” said magnitudes and through several codeepochs of said spreading code during said “B” time segments fordetermining “B” said magnitudes.
 11. The method of claim 1, wherein:integrating said incoming signal includes accumulating first “A” and “B”code correlations corresponding to said code phase increments duringfirst said “A” and first said “B” time segments for several epochs ofsaid spreading code; and continuing accumulating from said first “A” and“B” code correlations at said code phase increments during second said“A” and “B” time segments for several more epochs for providing “A” saidmagnitudes and “B” said magnitudes corresponding to said code phaseincrements.
 12. The method of claim 11, wherein: said continuingaccumulating includes inverting “A” and “B” code correlations duringsaid second “A” and “B” time segments.
 13. A receiver for acquiring anincoming signal having data bits spread by a spreading code, comprising:a correlation machine to integrate said incoming signal at code phaseincrements of said spreading code with two or more data bit searchpatterns in an “A” integration time period comprising at least twonon-contiguous “A” time segments for determining magnitudescorresponding to said code phase increments, respectively, for each ofsaid search patterns; the correlation machine further to integrate saidincoming signal at said code phase increments with said search patternsfor a “B” integration time period comprising “B” time segmentsalternating with said “A” time segments for determining magnitudescorresponding to said code phase increments, respectively, for each ofsaid search patterns; and a navigation signal processor configured touse a largest of said magnitudes for determining a particular one ofsaid code phase increments for receiving said incoming signal.
 14. Thereceiver of claim 13, further comprising: a data bit generator togenerate 2^((n−1)) said search patterns where said n is the number ofdata bit time lengths in a signal reception time length.
 15. Thereceiver of claim 13, further comprising: an AB timer to establish saidalternating “A” and “B” time segments for a signal reception time lengthof two or more said data bits; and wherein: the correlation machinefurther includes a depolarizer to depolarize a representation of saidincoming signal with said search patterns to provide “A” and “B” saidmagnitudes corresponding to said “A” and “B” integration time periods,respectively.
 16. The receiver of claim 13, further comprising: a databit generator to generate said search patterns including 00 or itsinverse and 01 or its inverse when a signal reception time length isabout two said data bit time lengths.
 17. The receiver of claim 13,further comprising: a data bit generator to generate said searchpatterns including 000 or its inverse, 001 or its inverse, 010 or itsinverse and 100 or its inverse when a signal reception time length isabout three data bit time lengths.
 18. The receiver of claim 13, furthercomprising: a data bit generator to generate a first of said searchpatterns having no inversion between a first and a second bit forproviding first “A” said magnitudes for said “A” integration time periodand first “B” said magnitudes for said “B” integration time period; anda second of said search patterns having an inversion between said firstand said second bit for providing second “A” said magnitudes for said“A” integration time period and second “B” said magnitudes for said “B”integration time period.
 19. The receiver of claim 13, furthercomprising: a data bit generator to generate a first of said searchpatterns having no inversion between a first and a second bit and noinversion between said second and a third bit for providing first “A”said magnitudes for said “A” integration time period and first “B” saidmagnitudes for said “B” integration time period; a second of said searchpatterns having no inversion between said first and said second bit andan inversion between said second and said third bit for providing second“A” said magnitudes for said “A” integration time period and second “B”said magnitudes for said “B” integration time period; a third of saidsearch patterns having an inversion between said first and said secondbit and an inversion between said second and said third bit forproviding third “A” said magnitudes for said “A” integration time periodand third “B” said magnitudes for said “B” integration time period; anda fourth of said search patterns having an inversion between said firstand said second bit and no inversion between said second and said thirdbit for providing fourth “A” said magnitudes for said “A” integrationtime period and fourth “B” said magnitudes for said “B” integration timeperiod.
 20. The receiver of claim 13, further comprising: an AB timer toestablish said time segments having time lengths about one-half a timelength of said data bits.
 21. The receiver of claim 13, wherein: saidincoming signal is a global navigation satellite system (GNSS) signal.22. The receiver of claim 13, wherein: the correlation machine isconfigured to perform code phase increment-by-increment codecorrelations between and said spreading code of said incoming signal anda local replica of said spreading code; and accumulating said codecorrelations through several code epochs of said spreading code duringsaid “A” time segments for determining “A” said magnitudes and throughseveral code epochs of said spreading code during said “B” time segmentsfor determining “B” said magnitudes.
 23. The receiver of claim 13,wherein: the correlation machine is configured to integrate saidincoming signal by accumulating first “A” and “B” code correlationscorresponding to said code phase increments during first said “A” and“B” time segments for several epochs of said spreading code; andcontinuing accumulating from said and “B” code correlations at said codephase increments during second said “A” and “B” time segments forseveral more epochs for providing “A” said magnitudes and “B” saidmagnitudes corresponding to said code phase increments.
 24. The receiverof claim 23, wherein: the correlation machine is further configured toinvert said “A” and “B” code correlations during said second “A” and “B”time segments.